Transconductance and current modulation for resonant frequency control and selection

ABSTRACT

In various embodiments, the invention provides a frequency controller and a temperature compensator for frequency control and selection in a clock generator and/or a timing and frequency reference. The various apparatus embodiments include a resonator adapted to provide a first signal having a resonant frequency; an amplifier; a temperature compensator adapted to modify the resonant frequency in response to temperature; and a process variation compensator adapted to modify the resonant frequency in response to fabrication process variation. In addition, the various embodiments may also include a frequency divider adapted to divide the first signal having the resonant frequency into a plurality of second signals having a corresponding plurality of frequencies substantially equal to or lower than the resonant frequency; and a frequency selector adapted to provide an output signal from the plurality of second signals. The output signal may be provided in any of various forms, such as differential or single-ended, and substantially square-wave or sinusoidal.

CROSS-REFERENCE TO RELATED APPLICATIONS

This application is related to, a conversion of and claims priority toU.S. Provisional Patent Application Ser. No. 60/555,193, filed Mar. 22,2004, inventor Michael Shannon McCorquodale, entitled “Monolithic andTop-Down Clock Synthesis with Micromachined Radio Frequency Reference”,which is commonly assigned herewith, the contents of which areincorporated herein by reference, and with priority claimed for allcommonly disclosed subject matter.

This application is related to and claims priority to U.S. patentapplication Ser. No. 11/084,962, filed concurrently herewith, inventorsMichael Shannon McCorquodale, Scott Michael Pernia, and Amar SarbbasebBasa, entitled “Monolithic Clock Generator and Timing/FrequencyReference”, which is commonly assigned herewith, the contents of whichare incorporated herein by reference, and with priority claimed for allcommonly disclosed subject matter.

FIELD OF THE INVENTION

The present invention, in general, relates to oscillation or clockingsignal generation, and more particularly, relates to a frequencycontroller and temperature compensator for a clock signal generator andtiming/frequency reference which is free-running, self-referenced,accurate over fabrication process, voltage and temperature, has lowjitter, and which may be monolithically integrated with other circuitryto form a single integrated circuit.

BACKGROUND OF THE INVENTION

Accurate clock generators or timing references have generally reliedupon crystal oscillators, such as quartz oscillators, which provide amechanical, resonant vibration at a particular frequency. The difficultywith such crystal oscillators is that they cannot be fabricated as partof the same integrated circuit (“IC”) driven by their clock signal. Forexample, microprocessors such as the Intel Pentium processor require aseparate clock IC. As a consequence, virtually every circuit requiringan accurate clock signal requires an off-chip clock generator.

There are several consequences for such non-integrated solutions. Forexample, because such a processor must be connected through outsidecircuitry (such as on a printed circuit board (PCB)), power dissipationis comparatively increased. In applications which rely on a finite powersupply, such as battery power in mobile communications, such additionalpower dissipation is detrimental.

In addition, such non-integrated solutions, by requiring an additionalIC, increase space and area requirements, whether on the PCB or withinthe finished product, which is also detrimental in mobile environments.Moreover, such additional components increase manufacturing andproduction costs, as an additional IC must be fabricated and assembledwith the primary circuitry (such as a microprocessor).

Other clock generators which have been produced as integrated circuitswith other circuits are generally not very accurate, particularly overfabrication process, voltage, and temperature (“PVT”) variations. Forexample, ring, relaxation and phase shift oscillators may provide aclock signal suitable for some low-sensitivity applications, but havebeen incapable of providing the higher accuracy required in moresophisticated electronics, such as in applications requiring significantprocessing capability. In addition, these clock generators oroscillators often exhibit considerable frequency drift, jitter, have acomparatively low Q-value, and are subject to other distortions fromnoise and other interference.

As a consequence, a need remains for a clock generator or timingreference which may be integrated monolithically with other circuitry,as a single IC, and which is highly accurate over PVT variations. Such aclock generator or timing reference should be free-running andself-referencing, and should not require locking or referencing toanother reference signal. Such as clock generator or timing referenceshould exhibit minimal frequency drift and have comparatively lowjitter, and should be suitable for applications requiring a highlyaccurate system clock. Such a clock generator or timing reference shouldalso provide multiple operating modes, including a clock mode, areference mode, a power conservation mode, and a pulsed mode.

SUMMARY OF THE INVENTION

In various exemplary embodiments, the invention provides a frequencycontroller and a temperature compensator for use in providing frequencycontrol and selection for a low-jitter, free-running andself-referencing clock generator and/or a timing and frequencyreference, which is highly accurate over PVT variations and which can beintegrated monolithically with other circuitry, to form a singularintegrated circuit. No separate reference oscillator is required. Thevarious exemplary embodiments of the invention include features forhighly accurate frequency generation over fabrication process, voltage,and temperature (“PVT”) variations. These features include frequencytuning and selection, and compensation for frequency variations whichmay be caused due to temperature and/or voltage fluctuations andfabrication process variations.

In addition, the various exemplary embodiments of the invention providea clock generator and/or a timing and frequency reference havingmultiple operating modes, including modes such as a power conservationmode, a clock mode, a reference mode, and a pulsed mode. In addition,the various embodiments provide multiple output signals at differentfrequencies, and provide low-latency and glitch-free switching betweenthese various signals.

Significantly, the various exemplary embodiments of the inventiongenerate a significantly and comparatively high frequency, such as inthe hundreds of MHz and GHz range, which is then divided to a pluralityof lower frequencies. Each such division by “N” (a rational number, as aratio of integers) results in a significant noise reduction, with phasenoise reduced by N and noise power reduced by N². As a consequence, thevarious exemplary embodiments of the invention result in significantlyless jitter than available with other oscillators, such as ringoscillators.

The various apparatus embodiments include a resonator, an amplifier, anda frequency controller, which may include various components or modulessuch as a temperature compensator, a process variation compensator, avoltage isolator, a frequency divider, and a frequency selector. Theresonator provides a first signal having a resonant frequency. Atemperature compensator adjusts the resonant frequency in response totemperature, and the process variation compensator adjusts the resonantfrequency in response to fabrication process variation. In addition, thevarious embodiments may also include a frequency divider to divide thefirst signal having the resonant frequency into a plurality of secondsignals having a corresponding plurality of frequencies substantiallyequal to or lower than the resonant frequency; and a frequency selectorto provide an output signal from the plurality of second signals. Thefrequency selector may further include a glitch-suppressor. The outputsignal may be provided in any of various forms, such as differential orsingle-ended, and substantially square-wave or sinusoidal.

The present invention provides an apparatus for frequency control of aresonator, the resonator adapted to provide a first signal having aresonant frequency, with the apparatus comprising an amplifiercoupleable to the resonator; and a frequency controller coupled to theamplifier and coupleable to the resonator, the frequency controlleradapted to modify the resonant frequency in response to at least onevariable of a plurality of variables. The plurality of variablescomprise temperature, fabrication process, voltage, and frequency, andthe amplifier may be a negative transconductance amplifier.

The frequency controller is further adapted to modify a current throughthe negative transconductance amplifier in response to temperature, andmay include a current source responsive to temperature. The currentsource may have one or more configurations selected from a plurality ofconfigurations, the plurality of configurations comprising CTAT, PTAT,and PTAT² configurations. The frequency controller may be furtheradapted to modify a current through or modify a transconductance of thenegative transconductance amplifier to select the resonant frequency.The frequency controller may be further adapted to modify a currentthrough the negative transconductance amplifier in response to a voltageor in response to fabrication process variation. The frequencycontroller may be further adapted to modify a transconductance of thenegative transconductance amplifier in response to fabrication processvariation.

In another exemplary embodiment, the present invention provides anapparatus, comprising a resonator adapted to provide a first signalhaving a resonant frequency; a negative transconductance amplifiercoupled to the resonator; and a temperature compensator coupled to thenegative transconductance amplifier and to the resonator, thetemperature compensator adapted to modify the resonant frequency inresponse to temperature.

The temperature compensator is further adapted to modify a currentthrough the negative transconductance amplifier in response totemperature, and may further comprise a current source responsive totemperature. The current source may have one or more configurationsselected from a plurality of configurations, the plurality ofconfigurations comprising CTAT, PTAT, and PTAT² configurations. Thecurrent source is typically coupled though one or more current mirrorsto the negative transconductance amplifier.

In another exemplary embodiment, the invention provides an apparatuscomprising a resonator, the resonator adapted to provide a first signalhaving a resonant frequency; a negative transconductance amplifiercoupled to the resonator; a current mirror coupled to the negativetransconductance amplifier; and a current source coupled to the currentmirror, the current source adapted to modify the resonant frequency byvarying a current through the current mirror and the negativetransconductance amplifier in response to temperature. The currentsource may have one or more configurations selected from a plurality ofconfigurations, the plurality of configurations comprising CTAT, PTAT,and PTAT² configurations.

The present invention may also include a mode selector coupled to thefrequency selector, wherein the mode selector is adapted to provide aplurality of operating modes, which may be selected from a groupcomprising a clock mode, a timing and frequency reference mode, a powerconservation mode, and a pulse mode.

For a reference mode, the invention may also include a synchronizationcircuit coupled to the mode selector; and a controlled oscillatorcoupled to the synchronization circuit and adapted to provide a thirdsignal; wherein in the timing and reference mode, the mode selector isfurther adapted to couple the output signal to the synchronizationcircuit to control timing and frequency of the third signal. Such asynchronization circuit may be a delay-locked loop, a phase-locked loop,or an injection locking circuit.

These and additional embodiments are discussed in greater detail below.Numerous other advantages and features of the present invention willbecome readily apparent from the following detailed description of theinvention and the embodiments thereof, from the claims and from theaccompanying drawings.

BRIEF DESCRIPTION OF THE DRAWINGS

The objects, features and advantages of the present invention will bemore readily appreciated upon reference to the following disclosure whenconsidered in conjunction with the accompanying drawings and exampleswhich form a portion of the specification, in which:

FIG. 1 (or “FIG. 1”) is a block diagram illustrating an exemplary systemembodiment in accordance with the teachings of the present invention.

FIG. 2 (or “FIG. 2”) is a block diagram illustrating a first exemplaryapparatus embodiment in accordance with the teachings of the presentinvention.

FIG. 3 (or “FIG. 3”) is a block diagram illustrating a second exemplaryapparatus embodiment in accordance with the teachings of the presentinvention.

FIG. 4 (or “FIG. 4”) is a high-level schematic and block diagramillustrating exemplary frequency controller, oscillator and frequencycalibration embodiments in accordance with the teachings of the presentinvention.

FIG. 5A (or “FIG. 5A”) is an exemplary graph illustrating oscillatorvoltage waveform (frequency) distortion with current injection into anoscillator.

FIG. 5B (or “FIG. 5B”) is an exemplary graph illustrating oscillatorvoltage waveform (frequency) distortion or variation with temperature.

FIG. 5C (or “FIG. 5C”) is an exemplary graph illustrating oscillatorfrequency as a function of the transconductance of a sustainingamplifier.

FIG. 6 (or “FIG. 6”) is a circuit diagram illustrating first exemplarynegative transconductance amplifier, temperature-responsive currentgenerator (I(T)), and LC tank oscillator embodiments in accordance withthe teachings of the present invention.

FIG. 7A (or “FIG. 7A”) is a circuit diagram illustrating an exemplarytemperature-responsive CTAT current generator in accordance with theteachings of the present invention.

FIG. 7B (or “FIG. 7B”) is a circuit diagram illustrating an exemplarytemperature-responsive PTAT current generator in accordance with theteachings of the present invention.

FIG. 7C (or “FIG. 7C”) is a circuit diagram illustrating an exemplarytemperature-responsive PTAT² current generator in accordance with theteachings of the present invention.

FIG. 7D (or “FIG. 7D”) is a circuit diagram illustrating an exemplarytemperature-responsive current generator, with selected CTAT, PTAT, andPTAT² configurations, in accordance with the teachings of the presentinvention.

FIG. 8 (or “FIG. 8”) is a circuit and block diagram illustrating secondexemplary negative transconductance amplifier, temperature-responsivecurrent generator (I(T)), and LC tank oscillator embodiments inaccordance with the teachings of the present invention.

FIG. 9 (or “FIG. 9”) is a circuit diagram illustrating an exemplarycontrolled capacitor module utilized in a frequency-temperaturecompensation module in accordance with the teachings of the presentinvention.

FIG. 10 (or “FIG. 10”) is a circuit diagram illustrating an exemplaryvoltage control module 650 utilized in a frequency-temperaturecompensation module in accordance with the teachings of the presentinvention.

FIG. 11 (or “FIG. 11”) is a circuit diagram illustrating an exemplaryfirst process variation compensation module in accordance with theteachings of the present invention.

FIG. 12 (or “FIG. 12”) is a circuit diagram illustrating an exemplarysecond process variation compensation module in accordance with theteachings of the present invention.

FIG. 13 (or “FIG. 13”) is a block diagram illustrating an exemplaryfrequency calibration module in accordance with the teachings of thepresent invention.

FIG. 14 (or “FIG. 14”) is a block diagram illustrating an exemplaryfrequency divider, square wave generator, asynchronous frequencyselector and glitch suppression module in accordance with the teachingsof the present invention.

FIG. 15 (or “FIG. 15”) is a graphical diagram illustrating exemplary lowlatency frequency switching in accordance with the teachings of thepresent invention.

FIG. 16 (or “FIG. 16”) is a block diagram illustrating an exemplaryfrequency divider in accordance with the teachings of the presentinvention.

FIG. 17 (or “FIG. 17”) is a block diagram illustrating an exemplarypower mode selection module in accordance with the teachings of thepresent invention.

FIG. 18 (or “FIG. 18”) is a block diagram illustrating an exemplarysynchronization module for a second oscillator in accordance with theteachings of the present invention.

FIG. 19 (or “FIG. 19”) is a flow diagram illustrating an exemplarymethod in accordance with the teachings of the present invention.

DETAILED DESCRIPTION OF THE EXEMPLARY EMBODIMENTS

While the present invention is susceptible of embodiment in manydifferent forms, there are shown in the drawings and will be describedherein in detail specific examples and embodiments thereof, with theunderstanding that the present disclosure is to be considered as anexemplification of the principles of the invention and is not intendedto limit the invention to the specific examples and embodimentsillustrated.

As indicated above, the various embodiments of the invention providenumerous advantages, including the ability to integrate a highlyaccurate (over PVT), low-jitter, free-running and self-referencing clockgenerator and/or a timing and frequency reference with other circuitry,such as illustrated in FIG. 1. FIG. 1 is a block diagram illustrating anexemplary system embodiment 150 in accordance with the teachings of thepresent invention. As illustrated in FIG. 1, the system 150 is a singleintegrated circuit, having a clock generator and/or timing/frequencyreference 100 of the present invention integrated monolithically withother, or second, circuitry 180, together with interface (I/F) (orinput/output (I/O) circuitry) 120. The interface 120 will generallyprovide power, such as from a power supply (not illustrated), ground,and other lines or busses to the clock generator 100, such as forcalibration and frequency selection. As illustrated, one or more outputclock signals are provided on bus 125, as a plurality of frequencies,such as a first frequency (f₀), a second frequency (f₁), and so on,through an (n+₁)^(th) frequency (f_(n)). In addition, a powerconservation mode (or low power mode (LP)) is provided (also on bus125). The second circuitry 180 (or the I/F 120) may also provide inputinto the clock generator 100, such as through selection signals (S₀, S₁,through S_(n)), and one or more calibration signals (C₀ through C_(n)).Alternatively, the selection signals (S₀, S₁, through S_(n)) and one ormore calibration signals (C₀ through C_(n)) may be provided directly tothe clock generator 100 through the interface 120, such as on bus 135,along with power (on line 140) and ground (on line 145).

The clock generator and/or timing/frequency reference 100, in additionto a low power mode, has additional modes discussed in greater detailbelow. For example, in a clock mode, the apparatus 100 will provide oneor more clock signals, as output signals, to the second circuitry 180.The second circuitry 180 may be any type or kind of circuitry, such as amicroprocessor, a digital signal processor (“DSP”), a radio-frequencycircuit, for example, or any other circuit which could utilize the oneor more output clock signals. Also for example, in a timing or frequencyreference mode, the output signal from the apparatus 100 may be areference signal, such as a reference signal for synchronization for asecond oscillator. As a consequence, the terminology clock generatorand/or timing/frequency reference will be utilized interchangeablyherein, with the understanding that the clock generator will alsogenerally provide a square-wave signal, which may or may not be providedwith a timing/frequency reference, which may utilize a substantiallysinusoidal signal instead. In addition, as discussed in greater detailbelow, the various embodiments of the invention also provided a pulsedmode, in which the output signal from clock generator and/ortiming/frequency reference 100 is provided in bursts or intervals, forincreased instruction processing efficiency and lower power consumption,for example.

It should be noted that the various signals are referred to as“substantially” sinusoidal or square-wave, for example. This is toaccommodate the various fluctuations, noise sources and otherdistortions introduced which may cause such signals to differ inpractice from the more ideal depictions found in textbooks. For example,as discussed in greater detail below, exemplary “substantially”square-wave signals are depicted in FIGS. 15A and 15B, and exhibit avariety of distortions, such as undershoots, overshoots, and othervariations, and are nonetheless considered to be very high qualitysquare-waves in practice.

Several important features of the present invention are in system 150.First, a highly accurate, low-jitter, free-running and self-referencingclock generator 100 is integrated monolithically with other (second)circuitry 180, to form a singular integrated circuit (system 150). Thisis in sharp contrast with the prior art, in which a reference oscillatoris used to provide a clock signal, such as a crystal referenceoscillator, which cannot be integrated with other circuitry and isoff-chip, as a second and separate device which must be connectedthrough a circuit board to any additional circuitry. For example, inaccordance with the present invention, the system 150, including clockgenerator 100, may be fabricated with other, second circuitry usingconventional CMOS, BJT, BiCMOS, or other fabrication technologiesutilized in modern IC manufacturing.

Second, no separate reference oscillator is required. Rather, inaccordance with the invention, the clock generator 100 isself-referencing and free-running, such that it is not referenced to orlocked to another signal, such as being synchronized in a phase lockedloop (“PLL”), delay locked loop (“DLL”), or via injection locking to areference signal, which is typical of the prior art.

Third, the clock generator 100 provides a plurality of outputfrequencies and a power conservation mode, such that frequencies may beswitched with low latency and in a glitch-free manner. For example,second circuitry 180 may shift to a power conservation mode, such as abattery or lower frequency mode, and request (through selection signals)a lower clock frequency for minimizing power consumption, or request alow power clock signal to enter a sleep mode. As discussed in greaterdetail below, such frequency switching is provided with substantiallynegligible latency, with low latency introduced for glitch prevention(in proportion to the number of glitch prevention stages utilized),using a merely a handful of clock cycles, rather than the thousands ofclock cycles required to change the output frequency from a PLL/DLLoscillator.

In addition, given the significantly high available output frequenciesof the clock generator and/or timing/frequency reference 100 discussedbelow, new operating modes are available. For example, clock start-uptimes are effectively or substantially negligible, allowing the clockgenerator and/or timing/frequency reference 100 to be repeatedly startedand stopped, such as turned off entirely or to be pulsed for powerconservation. For example, rather than running continuously as a clock,the clock generator and/or timing/frequency reference 100 can beoperated in comparatively short, discrete intervals or bursts (i.e.,pulsed), periodically or non-periodically, for instruction processing bya second circuit 180, such as a processor. As discussed in greaterdetail below, with the rapid start-up time, such pulsed operationprovides a power savings, as more instructions (million instructions persecond or MIPS) are processed per milliwatt (mW) of power consumption.In addition, such a pulsed mode may also be utilized to periodicallysynchronize a second clock or oscillator, in addition to other uses. Asa consequence, the clock generator and/or timing/frequency reference 100(and the other embodiments discussed below) has a plurality of operatingmodes, including a clock mode, a timing and/or frequency reference mode,a power conservation mode, and a pulsed mode.

Fourth, as discussed in greater detail below, the clock generator and/ortiming/frequency reference 100 includes features for highly accuratefrequency generation over fabrication process, voltage, and temperature(“PVT”) variations. These features include frequency tuning andselection, and compensation for frequency variations which may be causeddue to temperature and/or voltage fluctuations and fabrication processvariations.

Fifth, the clock generator and/or timing/frequency reference 100generates a significantly and comparatively high frequency, such as inthe hundreds of MHz and GHz range, which is then divided to a pluralityof lower frequencies. Each such division by “N” (a rational number, as aratio of integers) results in a significant noise reduction, with phasenoise reduced by N and noise power reduced by N². As a consequence, theclock generator of the present invention results in significantly lessjitter than available with other oscillators, such as ring oscillators.

These features are illustrated in greater detail in FIG. 2, which is ablock diagram illustrating a first exemplary apparatus 200 embodiment inaccordante with the teachings of the present invention. As illustratedin FIG. 2, the apparatus 200 is a clock generator and/ortiming/frequency reference, providing one or more output signals, suchas a clock or reference signal having any of a plurality of frequencies,selected using frequency selector 205. The apparatus (or clockgenerator) 200 includes an oscillator 210 (having a resonant element), afrequency controller 215, a frequency divider 220, a mode selector 225,and the frequency selector 205 mentioned above. In accordance with theinvention, the oscillator 210 generates a signal having a comparativelyhigh frequency, f₀. Due to PVT variations mentioned above, the frequencycontroller 215 is utilized to frequency select or tune the oscillator210, such that the oscillation frequency f₀ is selectable from aplurality of potential oscillation frequencies, i.e., the frequencycontroller 215 provides for output signals having frequencies which areaccurate over PVT variations.

For example, given these PVT variations, the output frequency from anoscillator, such as oscillator 210, may vary plus or minus 5%. For someapplications, such as those utilizing ring oscillators, such frequencyvariability may be acceptable. In accordance with the present invention,however, greater accuracy for the clock generator 200 is desirable,particularly for more sensitive or complex applications, such asproviding clock signals for integrated microprocessors,microcontrollers, digital signal processors, communication controllers,and so on. As a consequence, frequency controller 215 is utilized toadjust for these PVT variations, such that the output frequency from theoscillator is the selected or desired frequency f₀ with much lessvariance by several orders of magnitude, such as ±0.25% or less, andhaving a comparatively low-jitter.

To improve performance and decrease jitter (noise) and otherinterference, instead of generating a low frequency output andmultiplying it up to a higher frequency, as is typically done using PLLsand DLLs, the present invention generates a comparatively high frequencyoutput, f₀, which is then divided to one or more lower frequencies (f₁through f_(n)), using frequency divider 220. Clock signals having one ormore of the plurality of frequencies from frequency divider 220 may thenbe selected, using frequency selector 205. As indicated above, suchfrequency selection is provided glitch-free and with low latency,providing comparatively and significantly fast and glitch-free frequencyswitching. In addition, a plurality of operating modes are provided,using mode selector 225.

FIG. 3 is a block diagram illustrating in greater detail a secondexemplary apparatus embodiment, as clock generator and/ortiming/frequency reference 300, in accordance with the teachings of thepresent invention. Referring to FIG. 3, clock generator and/ortiming/frequency reference 300 comprises a resonator 310, a sustainingamplifier 305, a temperature compensator 315, a process variationcompensator 320, a frequency calibration module 325, one or morecoefficient registers 340, and depending on the selected embodiments,may also include a frequency divider and square wave generator 330, avoltage isolator 355, a resonant frequency selector 360, an outputfrequency selector 335 and mode selector 345. The sustaining amplifier305, temperature compensator 315, process variation compensator 320,voltage isolator 355, resonant frequency selector 360, and frequencycalibration module 325 are often included within a frequency controller,such as frequency controller 215. It should also be noted that thesquare-wave generator (of 330) may not be needed in timing or frequencyreference embodiments.

The resonator 310 may be any type of resonator which stores energy, suchas an inductor (L) and a capacitor (C) coupled to form an LC-tank, wherethe LC-tank has a selected configuration of a plurality of LC-tankconfigurations, or is otherwise electrically or electromechanicallyequivalent to or otherwise typically represented in the art as aninductor coupled to a capacitor. In addition to LC resonators, otherresonators are considered equivalent and within the scope of the presentinvention; for example, the resonator 310 may be a ceramic resonator, amechanical resonator (e.g., XTAL), a microelectromechanical (“MEMS”)resonator, or a film bulk acoustic resonator. In other cases, variousresonators may be represented by electrical or electromechanical analogyas LC resonators, and are also within the scope of the presentinvention. In exemplary embodiments, an LC-tank has been utilized as aresonator, to provide for a high Q-value.

The sustaining amplifier 305 provides for both start-up and sustainingamplification for the resonator 310. The temperature compensator 315provides frequency control for the resonator 310, to adjust theoscillation frequency based on variations due to temperature. Inselected embodiments, depending upon the degree of control desired orrequired, the temperature compensator 315 may include control over bothcurrent and frequency, as illustrated below for selected embodiments.Similarly, the process variation compensator 320 provides frequencycontrol for the resonator 310, to adjust the oscillation frequency basedon process variations inherent in semiconductor fabricationtechnologies, both process variations within a given foundry (e.g.,batch or run variations, variations within a given wafer, and die-to-dievariations within the same wafer) and process variations among differentfoundries and foundry processes (e.g., 130 nm and 90 nm processes).Frequency calibration module 325 is utilized to fine-tune and select thedesired output frequency, f₀, from among the oscillation frequencieswhich may occur in resonator 310, i.e., to select the output frequencyf₀ from a plurality of available or potential frequencies. In selectedembodiments, coefficient registers 340 are utilized to store coefficientvalues utilized in the various exemplary compensator and calibrationembodiments, discussed in greater detail below.

In addition to the temperature and process compensation, voltageisolator 355 provides isolation from variations in voltage, such as froma power supply, and may be implemented separately or as part of othercomponents, such as part of temperature compensator 315. In addition tofrequency adjustment for these PVT variations, the resonant frequencymay also be selected independently through resonant frequency selector360, for obtaining a selected frequency from a range of availablefrequencies.

For clock signal generation, clock generator 300 utilizes a frequencydivider (in module 330) to convert the output oscillation frequency f₀to a plurality of lower frequencies (f₁ through f_(n)) and to convert asubstantially sinusoidal oscillation signal to a substantially squarewave signal for clock applications, using a square wave generator (alsoin module 330). Frequency selector 335 then provides for selection ofone or more of the available output signals having the plurality offrequencies, and mode selector 345 may also provide for operating modeselection, such as providing a low power mode, a pulsed mode, areference mode, and so on. Using these components, the clock generator300 provides a plurality of highly accurate (over PVT), low jitter, andstable output frequencies, f₀, f₁ through f_(n), with minimal tonegligible frequency drift due to such PVT variations, thereby providingsufficient accuracy and stability for sensitive or complex applications,as mentioned above.

FIG. 4 is a high-level schematic and block diagram illustratingexemplary frequency controller, oscillator and frequency calibrationembodiments in accordance with the teachings of the present invention.As illustrated in FIG. 4, the resonator is embodied as a resonant LCtank 405, and the frequency controller is embodied as several elements,a negative transconductance amplifier 410 (used to implement thesustaining amplifier), a temperature-responsive (ortemperature-dependent) current generator (I(T)) 415, atemperature-responsive (or temperature-dependent) frequency (f₀(T))compensation module 420, a process variation compensation module 425,and may also include a frequency calibration module 430. The varioustemperature-responsive or temperature-dependent modules 415 and 420 aresensitive to or responsive to temperature fluctuations, and providecorresponding adjustments, such that the resonant frequency is accurateover these PVT variations.

The resonant LC tank 405 with a sustaining amplifier may be equallydescribed as a harmonic oscillator or harmonic core, and all suchvariations are within the scope of the present invention. It should benoted that while the resonant LC tank 405 is an inductor 435 in parallelwith a capacitor 440, other circuit topologies are also known andequivalent to that illustrated, such as an inductance in series with acapacitance. Another such equivalent topology is illustrated in FIG. 8.In addition, as indicated above, other types of resonators may beutilized and all are considered equivalent to the exemplary resonant LCtank illustrated herein. Moreover, as discussed in greater detail below,additional capacitances, both fixed and variable, are distributed in thevarious modules and effectively form part of the resonant LC tank 405.In addition, corresponding resistances (or impedances) R_(L) 445 andR_(C) 450 are illustrated separately, but should be understood to beintrinsic to the inductor 435 and capacitor 440, respectively, occurringas part of fabrication, and are not additional or separate componentsfrom the respective inductor 435 and capacitor 440. Conversely, suchresistances can also be included as part of compensation for PVTvariations.

The inductor 435 and capacitor 440 of the resonant LC tank or oscillator405 are sized to substantially or approximately provide the selectedoscillation frequency, f₀, or range of oscillation frequencies aroundf₀. In addition, inductor 435 and capacitor 440 may be sized to have orto meet IC layout area requirements, with higher frequencies requiringless area. Those of skill in the art will recognize that f₀≈½π√{squareroot over (LC)}, but only as a first order approximation because, asdiscussed below, other factors such as the impedances R_(L) and R_(C),along with temperature and process variations and other distortions,affect f₀. For example, the inductor 435 and capacitor 440 may be sizedto generate a resonant frequency in the 1–5 GHz range; in otherembodiments, higher or lower frequencies may be desirable, and all suchfrequencies are within the scope of the invention. In addition, theinductor 435 and capacitor 440 may be fabricated using any semiconductoror other circuitry process technology, and may be CMOS-compatible,bipolar-junction transistor-compatible, for example, while in otherembodiments, the inductor 435 and capacitor 440 may be fabricated usingsilicon-on-insulator (SOI), metal-insulator-metal (MiM),polysilicon-insulator-polysilicon (PiP), GaAs, strained-silicon,semiconductor heterojunction technologies, or MEMS-based(microelectromechanical) technologies, also for example and withoutlimitation. It should be understood that all such implementations andembodiments are within the scope of the invention. In addition, otherresonator and/or oscillator embodiments, in addition to or instead ofthe resonant LC tank 405, may also be utilized and are also within thescope of the present invention. As used herein, “LC tank” will mean andrefer to any and all inductor and capacitor circuit layouts,configurations or topologies which may provide oscillation, howeverembodied. It should be noted that the capability of the oscillator 405to be fabricated using a conventional process, such as CMOS technology,allows the clock generator to be fabricated integrally andmonolithically with other circuitry, such as the second circuitry 180,and provides a distinct advantage of the present invention.

In addition, the capacitance 440 illustrated in FIG. 4 is only a portionof the overall capacitance involved in the resonance and frequencydetermination of the resonant LC tank 405, and is a fixed capacitance.In selected embodiments, this fixed capacitance may representapproximately 10% to 90% of the total capacitance ultimately utilized inthe oscillator, as an example. As discussed in greater detail below, theoverall capacitance is distributed, such that additional fixed andvariable capacitance is selectively included within the clock generatorand/or timing/frequency reference 300, and is provided, for example, bytemperature-responsive frequency (f₀(T)) compensation module 420 andprocess variation compensation module 425, to provide for both selectingthe resonant frequency f₀ and to allow the resonant frequency f₀ to besubstantially independent of both temperature and process variations.

In the selected embodiments, the inductance 435 has been fixed, but alsocould be implemented in a variable manner, or as a combination of fixedand variable inductances. As a consequence, those of skill in the artwill recognize that the detailed discussions of fixed and variablecapacitance, for both frequency tuning and temperature and processindependence, pertain similarly to inductance choices. For example,different inductances could be switched in or out of the oscillator, tosimilarly provide tuning. In addition, a single inductor's inductancemay also be modulated. As a consequence, all such inductance andcapacitance variations are within the scope of the present invention.

Also as illustrated in FIG. 4, the resonant LC tank 405 and resultingoutput signal, referred to as a first (output) signal at nodes or lines470 and 475, is a differential signal and provides common-moderejection. Other configurations, including nan-differential or othersingle-ended configurations are also within the scope of the presentinvention. For example, in single-ended configurations, only oneinstantiation of the various modules (e.g., 485, 460) would be required,rather than the use of two for a balanced configuration as illustrated.Similarly, other components and features discussed below, such asfrequency dividers, would also have a single-ended rather thandifferential configuration. In addition, various embodiments illustratedutilize MOSFET transistors in various forms (such as CMOS, AMOS, IMOS,and so on); other implementations are also available, such as usingbipolar junction transistors (“BJTs”), BiCMOS, etc. All such embodimentsare considered equivalent and are within the scope of the presentinvention.

The negative transconductance amplifier 410 is selected to providetemperature compensation through transconductance (g_(m)) modulation andthe on-resistance of its resistors. Transconductance (g_(m)) modulationmay also be utilized independently in frequency selection. Anothersignificant advantage of the present invention is the selection of anegative transconductance amplifier 410 to provide start-up andsustaining amplification, because both oscillation amplitude andfrequency are affected by the transconductance of the sustainingamplifier, providing both amplitude modulation and frequency trimming(or tuning), in addition to providing temperature compensation. Thenegative transconductance amplifier 410 will inject current into theresonant LC tank 405 (and specifically onto the capacitor 440) inresponse to a voltage across the resonant LC tank 405, as illustrated(across nodes 470 and 475). That current injection, in turn, will change(and distort) the voltage waveform (as voltage is the integral of thecurrent), resulting in a change or variation in frequency, generally ininverse proportion to the magnitude of the transconductance, g_(m), asillustrated in FIG. 5A. It should be noted that this transconductance isa negative value, as gain is provided to cancel the loss intrinsic tothe resonant element. As a consequence, whenever “transconductanceamplifier” is utilized herein, it should be understood to mean and to bemerely an abbreviation for “negative transconductance amplifier”. Inturn, the transconductance is also a function of the bias current,substantially proportional (approximately) to the square root of thecurrent (yI(x)) through the amplifier 410 (for MOSFETs), andsubstantially proportional (approximately) to the current (yI(x))through the amplifier 410 (for BJTs), which is temperature-dependent,resulting in a waveform distortion which is both temperature andcurrent-bias dependent, as illustrated in FIG. 5B. In addition, asillustrated in FIG. 5C, the oscillation frequency is also related to anda function of the transconductance of the sustaining negativetransconductance amplifier 410, providing for oscillation frequencyselection. Moreover, in addition to temperature dependence (as I(T)),the current can also vary as a function of other variables (as I(x)),such as voltage or external tuning, may also be amplified such as by afactor of “y” (as illustrated below), and as a consequence, the currentis referred to as “yI(x)”.

Significant inventive breakthroughs of the present invention includeutilizing these potential distortions advantageously, to provide forfrequency compensation in generating the selected f₀ value of theoscillator, and frequency modulation through modulation of thetransconductance of the sustaining amplifier. As a consequence, and asdiscussed in greater detail below, the transconductance, first, may bemodified or varied for frequency selection, and second, to compensatefor such frequency variation due to temperature or voltage, by modifyingthe current yI(x), generally on a real-time or near real-time basis. Theselected frequency f₀, and its stability with respect to temperaturevariations, in accordance with the invention, may be determined throughappropriate selection of the transconductance g_(m) and selection ofI(T). Stated another way, in accordance with the present invention, thebias current is made temperature dependent, as I(T) (or, more generally,as yI(x)), which in turn affects the transconductance g_(m), which inturn affects the oscillation frequency f₀. This methodology may also beutilized for other variables, such as voltage fluctuations.

FIG. 6 is a circuit diagram illustrating exemplary negativetransconductance amplifier, temperature-responsive current generator(I(T)), and LC tank resonator embodiments in accordance with theteachings of the present invention. As illustrated in FIG. 6, theresonant LC tank 500 is coupled to a negative transconductance amplifierimplemented as a complementary cross-coupled pair amplifier 505(comprised of transistors M1, M2, M3 and M4) which, in turn, is coupledthrough a voltage isolator, implemented as current mirror 510(transistors 525A and 525B), to a temperature-responsive currentgenerator (I(x)) 515. The current mirror 510 may also be implemented ina cascode topology (520A and 520B), such as to provide improvedstability with variations in power supply and isolate the oscillatorfrom the power supply (voltage isolation). The temperature-responsivecurrent generator 515 may be implemented utilizing topologies such asCTAT (complementary to absolute temperature), PTAT (proportional toabsolute temperature), or PTAT² (proportional to absolute temperaturesquared), as illustrated in FIGS. 7A, 7B and 7C, respectively, andcombinations of CTAT, PTAT, and PTAT², as illustrated in FIG. 7D. Ineach case, the current I(T) (or yI(x)) injected into the negativetransconductance amplifier (complementary cross-coupled pair amplifier)505 has a temperature dependence, such as increasing current (PTAT andPTAT²) or decreasing current (CTAT) as a function of increasingtemperature, as illustrated. One or more combinations of thesetemperature-responsive current generators may also be implemented, asillustrated in FIG. 7D, such as CTAT in parallel with PTAT, for example.

The selection of a particular temperature-responsive ortemperature-dependent current generator is also a function of thefabrication process utilized; for example, CTAT may be utilized for aTaiwan Semiconductor (TSMC) fabrication process. More generally, asdifferent fabricators utilize different materials, such as aluminum orcopper, R_(L) typically varies, resulting in different temperaturecoefficients which, in turn, change the temperature coefficient of theoscillator, thereby requiring differences in I(T) compensation.Correspondingly, different ratios of CTAT, PTAT, and PTAT² compensationmay be required to provide an effectively flat frequency response as afunction of temperature. Not separately illustrated, the varioustemperature-responsive current generators illustrated in FIGS. 7A, 7B,7C and 7D may include a start-up circuit, which may be implemented asknown in the art. In addition, the transistors comprising the selectedtemperature-responsive current generator configuration may be biaseddifferently, such as biased in strong inversion for CTAT (M7 and M8) andPTAT² (M13 and M14), and in subthreshold for PTAT (M9 and M10) and PTAT²(M11 and M12), for the exemplary topologies illustrated.

FIG. 8 is a circuit and block diagram illustrating additional exemplarynegative transconductance amplifier, temperature-responsive (ortemperature-dependent) current generator (I(T) or I(x)), and LC tankoscillator embodiments in accordance with the teachings of the presentinvention. As illustrated in FIG. 8, the resonant LC tank 550 has adifferent topology than previously illustrated, but also is coupled to anegative transconductance amplifier implemented as a complementarycross-coupled pair amplifier 505 (transistors M1, M2, M3 and M4) which,in turn, is coupled through a plurality of current mirrors 510 (or 520)and 530 to a temperature-responsive (or temperature-dependent) currentgenerator (I(T) or I(x)) 515. As illustrated, the plurality of currentmirrors are utilized to successively provide gain to and increase thecurrent I(T) entering the negative transconductance amplifier 505 andresonant LC tank 550. Often, the tail device in the current mirror(e.g., transistor M6 in FIG. 6) providing current into node B and whichdrives the negative transconductance amplifier is selected to be a PMOSdevice, and thus several stages of mirroring may be required (as shown)to provide a PMOS current mirror input to the g_(m) amplifier. PMOS isoften selected because in modern CMOS processes, PMOS devices are oftenburied channel devices which are known to exhibit less flicker noisethan equally sized and similarly biased NMOS devices. Reduced flickernoise in the tail device reduces the phase noise and jitter of theoscillator because flicker noise is upconverted around the oscillationfrequency by the nonlinear active devices in the circuit.

As indicated above, the portion of the current mirror 510 or 520 (orother circuitry) sourcing current into the negative transconductanceamplifier 505 should have a high impedance at its output to reduce powersupply frequency drift, such as by using long transistor geometries andcascode configurations to increase output resistance, and providesignificant stability at node B. In addition, a shunt capacitor 570 alsomay be employed to filter and thereby reduce flicker noise from thevarious tail devices.

Depending upon the selected application, the use of the negativetransconductance amplifier 505 with its I(T) (or yI(x)) bias may providesufficient frequency stability, such that the additional frequencycontroller components may not be necessary or desirable in thatapplication. In other embodiments, however, additional accuracy and lessfrequency drift may be provided, using one or more of the componentsdiscussed in greater detail below.

In addition to providing a temperature-dependent current yI(x) (orI(T)), the various transistors M1, M2, M3 and M4 each have an associatedresistance during conduction, which may also tend to cause frequencydistortion and frequency drift during oscillation. In each half-cycle,either M1 and M4 or M2 and M3 are on and conducting. Such resistance isalso temperature dependent. As a consequence, the transistors M1, M2, M3and M4 should be adjusted in size (width and length) to also compensatefor such frequency effects. It should be noted that the current injectedinto the resonant LC tank 405 must be sufficient to sustain oscillation(as illustrated in FIG. 5C) and, as a consequence, will have a minimumvalue, which may limit the degree or capability of frequency controlwhich can be readily implemented through the negative transconductanceamplifier 410 (or 505) and temperature-dependent current generator 415(or 515). As a consequence, I(T) and the transistor (M1, M2, M3 and M4)sizing should be jointly selected to provide for oscillation start up,to accommodate maximum currents for power consumption constraints, andto fit into the selected IC area and layout. For example, thetransconductance g_(m) may be selected to provide approximatelysufficient current to ensure start up and sustain oscillation, with afrequency characteristic of decreasing frequency with increasingtemperature, followed by sizing transistors M1, M2, M3 and M4 to belarge enough to either make the frequency independent of temperature orincreasing with increasing temperature, followed by fine-tuning thefrequency-temperature relationship with appropriate selection of I(T).In selected modeled embodiments, this has resulted in frequency accuracyof approximately ±0.25% to 0.5% over PVT, which may be more thansufficient for many applications.

Referring again to FIG. 4, additional compensation modules are alsoutilized to provide greater control and accuracy over the resonantfrequency f₀, such as for applications in which greater accuracy andless variance (or frequency drift) may be required, or wheretechnologies do not allow the previous techniques to provide sufficientaccuracy over PVT variations, such as to provide a frequency accuracy ofapproximately ±0.25% or better. In these circumstances,temperature-dependent (or temperature-responsive) frequency (f₀(T))compensation module 420 may be utilized, such as the exemplarytemperature-responsive frequency (f₀(T)) compensation module 420. Thismodule 420 may be implemented, for example, utilizing controllablecapacitance modules 485, with each coupled to a respective side or railof the resonant LC tank 405 (lines 470 and 475), and with each undercommon control, provided by a first plurality (“w”) of switchingcoefficients (p₀ though P_((w−1))) (register 495) and a voltagecontroller (V_(CTLR)) 480 providing a control voltage determined by asecond plurality (“x”) of switching coefficients (q₀ thoughq_((x−1)(register 455), with representative examples illustrated inFIGS. 9 and 10.

FIG. 9 is a circuit diagram illustrating an exemplary controllablecapacitance module 635 in accordance with the teachings of the presentinvention, which may be utilized as the controllable capacitance modules485 in the frequency-temperature compensation module 420 (and attachedto each side of the resonant LC tank 405 (nodes or lines 470 and 475)).As illustrated, the controllable capacitance module 635 is comprised ofa bank or array of a plurality (w) of switchable capacitive modules 640of binary-weighted fixed capacitors (C_(f)) 620 and variable capacitors(varactors) (C_(v)) 615. Any type of fixed capacitors 620 and variablecapacitors (varactors) 615 may be utilized; in selected embodiments, thevaractors 615 are A-MOS (accumulation mode MOSFET), I-MOS (inversionmode MOSFET), and/or junction/diode varactors. Each switchablecapacitive module 640 has an identical circuit layout, and each differsby a binary weighted capacitance, with switchable capacitive module 640₀ having a capacitance of one unit, switchable capacitive module 64O₁having a capacitance of two units, and so on, with switchable capacitivemodule 640 _((w−1)) having a capacitance of 2^((w−1)) units, with eachunit representing a particular capacitance value (typically infemtofarads (fF) or picofarads (pF)).

Within each switchable module 640, each fixed and variable capacitanceis initially equal, with the variable capacitance allowed to vary inresponse to the control voltage provided at node 625. This controlvoltage, in turn, varies with temperature, resulting in an overall ortotal capacitance provided by the controlled capacitor module 635 alsovarying as a function of temperature and which, in turn, is utilized tovary the resonant frequency f₀. Also within each switchable capacitivemodule 640, either the fixed capacitance C_(f) or the variablecapacitance C_(v) is switched into the circuit, not both, usingswitching coefficients p₀ though p_((w−1)). For example, in the selectedembodiment, for a given or selected module 640, when its corresponding“p” coefficient is a logic high (or high voltage), the correspondingfixed capacitance C_(f) is switched into the circuit and thecorresponding variable capacitance C_(v) is switched out of the circuit(and coupled to a power rail V_(DD) or ground (GND), depending whetherthe device is AMOS or IMOS, respectively, to avoid a floating node andto minimize the capacitance presented to the tank), and when itscorresponding “p” coefficient is a logic low (or low voltage), thecorresponding fixed capacitance C_(f) is switched out of the circuit andthe corresponding variable capacitance C_(v) is switched into thecircuit and coupled to the control voltage provided on node 625.

In an exemplary embodiment, a total of eight switchable capacitivemodule 640 (and corresponding first plurality of eight switchingcoefficients p₀ though p₇ have been implemented to provide 256combinations of fixed and variable capacitances. As a result,significant control over oscillation frequency as a function oftemperature variations is provided.

FIG. 10 is a circuit diagram illustrating an exemplary temperaturedependent voltage control module 650 utilized to provide the controlvoltage in the controllable capacitance module 635 (of thefrequency-temperature compensation module 420) in accordance with theteachings of the present invention. As illustrated, voltage controlmodule 650 creates a temperature-dependent current I(T) (or moregenerally, a current I(x)), using current generator 655, as previouslydiscussed, using one or more combinations of PTAT, PTAT² and/or CTATcurrent generators, and may share the I(T) generator 415 utilized withthe negative transconductance amplifier 410, instead of providing aseparate generator 655. The temperature-dependent current I(T) (or I(x))is mirrored through current mirror 670 to an array or bank of aplurality of switchable resistive modules or branches 675 and a fixedcapacitive module or branch 680, all configured in parallel. Theresistors 685 may be any type or combination of different types, such asdiffusion resistors (p or n), polysilicon, metal resistors, salicide orunsalicide polysilicon resistors, or well resistors (p or n well), forexample. Each switchable resistive module 675 is switched in or out ofthe voltage control module 650 by a corresponding “q” coefficient of asecond plurality (“x”) of switching coefficients q₀ though q_((x−1)).When switchable resistive module 675 is switched into the circuit (suchas when its corresponding coefficient is a logic high or high voltage),the resulting voltage across its corresponding resistor 685 is alsotemperature-dependent, due to the temperature-dependent current I(T). Ina selected embodiment, three switchable resistive modules 675 wereutilized, providing 8 branch combinations. As a result, the controlvoltage provided to node 625 is also a function of temperature, therebyproviding a temperature dependence or sensitivity to the variablecapacitors 615 in controllable capacitance module 635.

The first of switching coefficients p₀ through p_((w−1)) and the secondplurality of switching coeffficients q₀ though q(_(x−1)) are determinedpost-fabrication by testing a representative IC having the clockgenerator of the present invention. In the exemplary embodiments, thefirst plurality of switching coefficients p₀ through p_((w−1)) aredetermined first, by testing various combinations of coefficients, toprovide a coarse level of adjustment, resulting is a substantially ofmostly flat frequency response as a function of varying ambienttemperature. The second plurality of switching coefficients q₀ thoughq_((x−1)) are then determined, also by testing various combinations ofcoefficients, to provide a finer level of adjustment, resulting is asubstantially flat frequency response as a function of varying ambienttemperature. The first and second plurality of coefficients are thenloaded into respective registers 495 and 455 in all of the ICsfabricated in the selected processing run (or batch). Depending id thefabrication processing, under other circumstances, it is possible thatfor higher accuracy, each IC may be separately calibrated. As result, inconjunction with the temperature compensation provided by the negativetransconductance amplifier 410 and I(T) generator 415, the overallfrequency response of the clock generator is substantially independentof temperature fluctuations.

As a consequence, the overall capacitance provided to the resonant LCtank 405 is distributed into a combination of fixed and variableportions, with the variable portions responsive to provide temperaturecompensation and, therefore, control over the resonant frequency f₀. Themore variable capacitance C_(v) which is switched into the circuit(controlled capacitor module 635), the greater the response tofluctuations in ambient temperature.

In addition to providing temperature compensation, it should be notedthat a switched or controllable capacitance module 635 may also beutilized to select or tune the resonant frequency f₀.

Referring again to FIG. 4, another compensation module is also utilizedto provide greater control and accuracy over the resonant frequency f₀,also for applications in which greater accuracy and less variance (orfrequency drift) may be required, such as to provide a frequencyaccuracy of approximately ±0.25% or better over PVT. In thesecircumstances, a process variation compensation module 425 may beutilized, to provide control over the resonant frequency f₀independently of fabrication process variations, such as the exemplarymodules illustrated in FIGS. 11 and 12.

FIG. 11 is a circuit diagram illustrating an exemplary first processvariation compensation module 760 in accordance with the teachings ofthe present invention. The first process variation compensation module760 may be utilized as the process compensation modules 460 in FIG. 4,with each module attached to a rail or side of the resonant LC tank 405(lines 470 and 475). In addition, each first process variationcompensation module 760 is controlled by a third plurality (“y”) ofswitching coefficients r₀ though r_((y−1)), stored in register 465. Thefirst process variation compensation module 760 provides an array ofswitchable capacitive modules having binary-weighted, first fixedcapacitances 750, for adjustment and selection of the resonant frequencyf₀, by switching in or out a plurality of fixed capacitances 750,through a corresponding plurality of switching transistors 740(controlled by a corresponding “r” coefficient).

Again, as each capacitance branch is switched in or out of the array orcircuit 760, the corresponding first fixed capacitance is added orsubtracted from the total capacitance available for oscillation in theresonant LC tank, thereby modulating the resonant frequency. The thirdplurality of switching coefficients r₀ though r_((y−1)) is alsodetermined post-fabrication using test ICs, generally as an iterativeprocess with the determinations of the first and second pluralities ofswitching coefficients. This calibration is accomplished using thefrequency calibration module (325 or 430) and a reference oscillatorknown to have a predetermined frequency. The determined “r” coefficientsare then stored in the corresponding registers 465 of the ICs of thatproduction or process batch. Alternatively, each IC may be calibratedseparately, for example.

To avoid additional frequency distortions, several additional featuresmay be implemented with this first process variation compensation module760. First, to avoid additional frequency distortion, the on resistanceof the MOS transistors 740 should be small, and therefore thetransistors' width/length ratio is large. Second, large capacitances maybe split into two branches, with two corresponding transistors 740controlled by the same “r” coefficient. Third, to provide for theresonant LC tank to have a similar load under all conditions, when afirst fixed capacitance 750 is switched in or out of the circuit 760, acorresponding second fixed capacitance 720, as a “dummy” capacitor(having a significantly smaller capacitance or the smallest size allowedby the design rules for the fabrication process), is correspondinglyswitched out of or into the circuit, based on the inverse of thecorresponding “r” coefficient. As a consequence, approximately orsubstantially the same on resistance of the transistors 740 is alwayspresent, with only the amount of capacitance varied.

As an alternative to the use of the “dummy” capacitances, metal fuses orthe like could be utilized instead of the transistors 740. Metal fuseswould be left intact to include the corresponding fixed capacitance 750,and could be “blown” (open-circuited) to remove the corresponding fixedcapacitance 750 from the resonant LC tank 405.

FIG. 12 is a circuit diagram illustrating an exemplary second processvariation compensation module 860 in accordance with the teachings ofthe present invention. The second process variation compensation module860 may be utilized as the process compensation modules 460 in FIG. 4,with each module attached to a rail or side (lines 470 and 475) of theresonant LC tank 405, instead of modules 760. In addition, each secondprocess variation compensation module 760 would also be controlled by athird plurality of switching coefficients r₀ though r_((y−1)), stored inregister 465. (Because of the different circuitry employed in eachexemplary process variation compensation module 760 or 860, however, thecorresponding third pluralities of switching coefficients r₀ thoughr_((y−1)) would, of course, be different from each other.)

It should be noted that FIG. 12 provides a varactor illustrationdifferent from those utilized in other Figures, in which a varactor 850is represented by a MOS transistor, rather than as a capacitor with anarrow through it. Those of skill in the art will recognize thatvaractors are often A-MOS or I-MOS transistors, or more generally MOStransistors, such as those illustrated in FIG. 12, and configured byshorting the transistor's source and drain. As a consequence, the otherillustrated varactors may be consider to include, as potentialembodiments, the A-MOS or I-MOS transistors as configured as in FIG. 12.In addition, the varactors 850 are also binary-weighted with respect toeach other.

The second process variation compensation module 860 has a similarstructural concept, but additional notable differences from the firstprocess variation compensation module 760. The second process variationcompensation module 860 provides an array or bank of a plurality ofswitchable variable capacitive modules 865, without MOSswitches/transistors, and hence the losses or loading through the MOStransistors are eliminated. Instead, the load appears as a low losscapacitance; such low loss also implies that the oscillator start-uppower is less. In the second process variation compensation module 860,a MOS varactor 850 is switched either to ground or the power rail(voltage V_(DD)), thereby providing either the minimum capacitance orthe maximum capacitance to the resonant LC tank 405 based upon thevaractor 850 geometry. For AMOS, switched to voltage V_(DD) wouldprovide minimum capacitance and switched to ground would provide maximumcapacitance, while the opposite is the case for IMOS. Again, the secondprocess variation compensation module 860 is comprised of an array ofbinary-weighted variable capacitances, as varactors 850, for adjustmentand selection of the resonant frequency f₀, by coupling a selectedvaractor 850 to ground or V_(DD), through a corresponding “r”coefficient.

As each capacitance branch is switched to ground or V_(DD), thecorresponding variable capacitance is added to or not included in thetotal capacitance available for oscillation in the resonant LC tank,thereby modulating the resonant frequency. More particularly, for anA-MOS implementation, coupling to V_(DD) (as V_(in)) provides lessercapacitance and coupling to ground (V_(in)=0) provides greatercapacitance, with the opposite holding for an I-MOS implementation, inwhich coupling to V_(DD) (as V_(in)) provides greater capacitance andcoupling to ground (V_(in)=0) provides lesser capacitance, where it isassumed that the voltage on the rails of the LC tank (nodes or lines 470and 475 of FIG. 4) is between zero V and voltage V_(DD), andsignificantly or substantially far from either voltage level. The thirdplurality of switching coefficients r₀ though r_((y−1)) is alsodetermined post-fabrication using test ICs, also generally as aniterative process with the determinations of the first and secondpluralities of switching coefficients. The determined “r” coefficientsare then stored in the corresponding registers 465 of the ICs of thatproduction or process batch. Again, individual ICs may also becalibrated and tested separately.

It should also be noted that the illustrated embodiments for modulessuch as temperature compensator 315 (or 410 and 415) and processvariation compensator 320 (or 425 and 460), such as those illustrated inFIGS. 6–12, may be utilized for other purposes. For example, the variousillustrated embodiments for the compensator 315 (or 410 and 415) may bemade dependent upon process variation, rather than temperature.Similarly, the various illustrated embodiments for the compensator 320(or 425 and 460) may be made dependent upon temperature, rather thanprocess variation. As a consequence, the embodiments for these and othermodules should not be considered limited to the exemplary circuits andstructures illustrated, as those of skill in the art will recognizeadditional and equivalent circuits and applications, all of which arewithin the scope of the invention.

Referring again to FIGS. 3 and 4, the clock generator and/ortiming/frequency reference 300 may also include a frequency calibrationmodule (325 or 430). This frequency calibration module is the subject ofa separate patent application, but its high-level functionality isdescribed briefly below. FIG. 13 is a high-level block diagramillustrating an exemplary frequency calibration module 900 (which may beutilized as module 325 or 430) in accordance with the teachings of thepresent invention. The frequency calibration module 900 includes adigital frequency divider 910, a counter-based frequency detector 915, adigital pulse counter 905, and a calibration register 930 (which alsomay be utilized as register 465). Using a test IC, the output signalfrom the clock generator (200 or 300) is frequency divided (910) andcompared with a known reference frequency 920 in frequency detector 915.Depending upon whether the clock generator (200 or 300) is fast or slowwith respect to the reference, down or up pulses are provided to thepulse counter 905. Based upon those results, the third plurality ofswitching coefficients r₀ though r_((y−l)) is determined, and the clockgenerator (200 or 300) is calibrated to a selected reference frequency.Again, individual ICs may also be calibrated and tested separately.

Referring again to FIGS. 2, 3 and 4, it will be appreciated by those ofskill in the art that a highly accurate over PVT, low jitter,free-running and self-referenced oscillator has been described,providing a differential, substantially sinusoidal signal having aselectable and tunable resonant frequency, f₀, available at nodes 470and 475. For many applications, this signal is sufficient, and may beutilized directly (and may be output on line 250 of FIG. 2, or line 350of FIG. 3, or between the rails or lines 470 and 475 of FIG. 4). Forexample, this signal may be utilized as a timing or frequency reference.In accordance with the present invention, additional applications areavailable, including clock generation (substantially square wave),frequency division, low-latency frequency switching, and mode selection,as described below.

FIG. 14 is a block diagram illustrating an exemplary frequency dividerand square wave generator 1000, and an exemplary asynchronous frequencyselector 1050, with exemplary glitch suppression module 1080 inaccordance with the teachings of the present invention. As indicatedabove, frequency divider and square wave generator 1000 may be includedin or comprise modules 220 and/or 330, and frequency selector 1050 (withor without glitch suppression module 1080) may be included in orcomprise modules 205 and/or 335.

Referring to FIG. 14, the output signal from the oscillator, namely, asubstantially sinusoidal signal having a frequency f₀, such as output online 250 of FIG. 2, or line 350 of FIG. 3, or between the rails or lines470 and 475 of FIG. 4, is input into frequency divider and square wavegenerator 1000. The frequency of this substantially sinusoidal signal isdivided by any one or more arbitrary values “N” into “m” differentfrequencies (including f₀, where appropriate), and converted tosubstantially square wave signals, resulting in a plurality ofsubstantially square wave signals having m+1 different availablefrequencies, output on lines or bus 1020 as frequencies f₀, f₁, f₂,through f_(m). Any of these substantially square wave signals having m+1different available frequencies are selectable asynchronously throughexemplary asynchronous frequency selector 1050 which, as illustrated,may be embodied as a multiplexer. The selection of any of thesesubstantially square wave signals having m+1 different availablefrequencies may be accomplished through the plurality of selection lines(S_(m) through S₀) 1055, providing a substantially square wave signalhaving the selected frequency, output on line 1060.

As part of asynchronous frequency selection, glitch suppression is alsoprovided by glitch suppression module 1080, which may be embodied in aplurality of ways, including through the use of one or more exemplary Dflip-flops (“DFFs”) illustrated in FIG. 14. A glitch could occur in anasynchronous frequency transition in which either a low state or a highstate is not maintained for a sufficient period of time and may causemetastability in circuitry which is driven by the output clock signal.For example, an asynchronous frequency transition could result in a lowstate at a first frequency transitioning into a high state at a secondfrequency, at a point where the high state is about to transition backto a low state at the second frequency, resulting in a voltage spike orglitch. To avoid potential glitches from being provided as part of anoutput clock signal, the selected substantially square wave signal(having the selected frequency) is provided on line 1060 to a first DFF1065 which provides a holding state; if a glitch should occur, it willbe held until a clock edge triggering the DFF. To avoid the glitchoccurring at the clock edge, the DFFs may be clocked at less than themaximum available frequency, or one or more additional DFFs (such as DFF1070) may be employed, as during the wait for another clock signal, theQ output from the DFF 1065 will have stabilized to either a first state(high or low) or a second state (low or high), such as to either thepower or ground rail. It has been shown by the inventors that 2 DFFs aresufficient, with additional DFFs potentially being added as may bedesired, but with additional DFFs causing increased switching latency.While illustrated utilizing exemplary DFFs, other flip-flops or countersmay be utilized, and those of skill in the art will recognize myriadother equivalent implementations which will achieve this result, and allsuch variations are within the scope of the invention.

Such exemplary low latency frequency switching in accordance with theteachings of the present invention is illustrated in FIG. 15. FIG. 15 isalso illustrative of “substantially” square waves of the presentinvention, which are typical of actual square waves utilized in varioustechnologies, exhibiting reasonable variation, undershoots andovershoots at their respective high and low states (and not the perfect“flatness” of textbook examples). FIG. 15, part A, illustratesasynchronous glitch-free switching from 1 MHz to 33 MHz, while part Billustrates measured glitch-free switching from 4 MHz to 8 MHz, then to16 MHz, and then to 33 MHz.

Referring again to FIG. 14, the frequency divider and square wavegenerator 1000 may be implemented in innumerable ways, such asdifferential or single-ended, with the illustrated divider being merelyexemplary. As the output from the oscillator illustrated in FIG. 4 isdifferential (across lines or rails 470 and 475), the first divider 1005is also differential and provides complementary outputs, to present asubstantially constant load to the oscillator and to maintain phasealignment, and is fast, to support high frequencies such as in the GHzrange. In addition, it may be necessary or advisable to reject anyrelaxation mode oscillation of the first divider 1005. The seconddivider 1010 may also be differential and provide any arbitraryfrequency division (divide by “M”), such as dividing by an integer, amultiple of two, a rational number, or any other amount or number, etc.Topologies or configuration for such dividers are known in the art, andany such divider may be utilized. Such dividers, for example and withoutlimitation, may be a sequence (multiple stages) of counters orflip-flops 1075, such as those illustrated in FIG. 16, which providefrequency division in powers or multiples of 2, with the output of eachstage providing a clock signal for the next stage and also fed back toits own input, as illustrated. As illustrated, a plurality offrequencies are then available for output on lines or bus 1020, such asf₀/2, f₀/4, and so on, through f₀/2^(N). In addition, as illustrated,buffers 1085 may also be utilized, from the oscillator to the firstdivider 1005, to provide sufficient voltage to drive the divider 1005,and also between divider stages, to isolate state-dependent loadvariation which could also affect signal rise and fall times.

It should also be noted that the use of the various flip-flops has alsoprovided a substantially square wave, as any substantially sinusoidalsignal has been provided to clock a flip flop, whose output is thenpulled to a high or low voltage. Other square wave generators may alsobe utilized, as known or becomes known in the art. In the illustratedembodiments, to maintain phase alignment, differential signals aremaintained through the last division. Following the last frequencydivision, the plurality of signals (each having a different frequency)are then squared (in module 1015) to provide substantially an evenlydivided (e.g. 50:50) duty cycle, such that the time in which the signalis in a first (high) state is substantially equal to the time in whichthe signal is in a second (low) state.

FIG. 17 is a block diagram illustrating an exemplary mode selectionmodule in accordance with the teachings of the present invention. Thereare circumstances in which a highly-accurate, high performancereference, such as a clock generator (100, 200 or 300) of the invention,is unnecessary, such as in a low power, standby mode. In thesecircumstances, in accordance with the invention, either no clock outputis provided, or a low power, reduced performance clock 1105 output isprovided. For example, at comparatively low frequencies, a lowperformance ring oscillator may provide suitable performance with lowpower consumption. As illustrated in FIG. 17, for these conditions, theoutput of the low power oscillator 1105 may be selected (throughmultiplexer 1100), and provided as a clock output to other circuitry. Athigher frequencies, however, such low performance oscillators consumeconsiderably more power, typically significantly more than theoscillator of the present invention. There is typically a “break-even”point as a function of frequency, after which the clock generator (100,200 or 300) provides both higher performance and lower powerconsumption, and may be selected (through multiplexer 1100), andprovided as a clock output to other circuitry. As a consequence, theclock generator (100, 200 or 300) may also be utilized to provide a lowpower mode.

In addition, using mode selector 1110, other modes may be selected, suchas a no power mode, rather than merely a low-frequency or sleep mode, asthe clock generator (100, 200 or 300) may be restarted comparativelyrapidly, or a pulsed mode, in which the clock generator (100, 200 or300) is repeatedly stopped and restarted, periodically ornon-periodically, in bursts or intervals. Various reference modes arediscussed below.

In sharp contrast to the prior art, this pulsed clocking using the clockgenerator and/or timing/frequency reference (100, 200 or 300) of thepresent invention provides power savings or conservation. While morepower may be consumed during a given burst, as the clock has acomparatively high frequency, more instructions are processed in thatinterval, followed by no or limited power dissipation during thenon-pulse or off interval, resulting in higher MIPS/mW compared to acontinuously running clock. In contrast, due to the comparatively longstart-up time and locking of prior art clocks, such pulsed clockingresults in more power consumption and less efficiency in the prior art.

FIG. 18 is a block diagram illustrating an exemplary synchronizationmodule 1200 for a second oscillator in accordance with the teachings ofthe present invention. As mentioned above, the clock generator and/ortiming/frequency reference (100, 200 or 300) may provide a referencemode to synchronize other oscillators or clocks, which may or may not below power, such as second oscillator 1210 (e.g., ring, relaxation, orphase shift oscillators). An output signal from the clock generatorand/or timing/frequency reference (100, 200 or 300) is further frequencydivided as needed to form a plurality of available referencefrequencies, with a reference frequency selected from this plurality offrequencies. This may be accomplished using the modules discussed above,such as by using the existing frequency dividers (220, 330, 1000, forexample), and then providing the reference signal from the frequencyselector 1050 (or 205 or 335). For example, referring to FIG. 3, modeselector 345 may select a reference mode and provide the outputreference signal from frequency selector 335 to a second oscillator(with synchronization module) 375. A synchronization module, such as PLLor DLL 1205, is then utilized to synchronize the output signal from thesecond oscillator 1210 to the reference signal provided by clockgenerator and/or timing/frequency reference (100, 200 or 300). Inaddition to a mode of continuous synchronization, apulsed-synchronization may also be provided, in which the clockgenerator and/or timing/frequency reference (100, 200 or 300) provides apulsed output, and synchronization occurs during the interval of thesepulses, as a synchronization interval.

FIG. 19 is a flow diagram illustrating an exemplary method in accordancewith the teachings of the present invention, and provides a usefulsummary. The method begins with start step 1220, such as through clockgenerator and/or timing/frequency reference (100, 200 or 300) start-up.It should be noted that while illustrated in FIG. 19 as consecutivesteps, these steps may occur in any order, and generally may occurconcurrently as the clock generator and/or timing/frequency reference(100, 200 or 300) operates. Referring to FIG. 19, a resonant signalhaving a resonant frequency is generated, step 1225, such as through LCtank 405 or resonator 310. The resonant frequency is adjusted inresponse to temperature, step 1230, such as through a temperaturecompensator 315, which adjusts current and frequency. The resonantfrequency is adjusted in response to fabrication process variation, step1235, such as through process variation compensator 320. The resonantsignal having the resonant frequency is divided into a plurality ofsecond signals having a corresponding plurality of frequencies, in whichthe plurality of frequencies are substantially equal to or lower thanthe resonant frequency, step 1240, such as through frequency divider 330or 1000). An output signal is selected from the plurality of secondsignals, step 1245, such as through frequency selector 335 or 1050, forexample. Depending upon the selected embodiment or mode, the selectedoutput signal may be provided directly, for example, as a referencesignal.

In other embodiments, such as when the output signal is a differentialrather than single-ended signal, and when the resonant signal is asubstantially sinusoidal signal, the method continues with convertingthe differential, substantially sinusoidal signal to a single-ended,substantially square wave signal having a substantially equal high andlow duty cycle, as needed, step 1250, such as to generate a clock outputsignal using modules 330 or 1000, for example. An operating mode is alsoselected from a plurality of operating modes, step 1255, where theplurality of operating modes can be selected from a group comprising aclock mode, a timing and frequency reference mode, a power conservationmode, and a pulse mode, for example, such as using mode selector 225 or345. When a reference mode is selected in step 1255, in step 1260, themethod proceeds to step 1265, to synchronize a third signal (e.g., froma second oscillator) in response to the output signal, such asillustrated in FIG. 18. Following steps 1260 or 1265, the method may endor repeat (continue) (such as with the clock generator and/ortiming/frequency reference (100, 200 or 300) running continuously),return step 1270.

Also in summary, the present invention provides an apparatus forfrequency control of a resonator, the resonator adapted to provide afirst signal having a resonant frequency, with the apparatus comprisingan amplifier coupleable to the resonator; and a frequency controllercoupled to the amplifier and coupleable to the resonator, the frequencycontroller adapted to modify the resonant frequency in response to atleast one variable of a plurality of variables. The plurality ofvariables comprise temperature, fabrication process, voltage, andfrequency, and the amplifier may be a negative transconductanceamplifier.

The frequency controller is further adapted to modify a current throughthe negative transconductance amplifier in response to temperature, andmay include a current source responsive to temperature. The currentsource may have one or more configurations selected from a plurality ofconfigurations, the plurality of configurations comprising CTAT, PTAT,and PTAT² configurations. The frequency controller may be furtheradapted to modify a current through or modify a transconductance of thenegative transconductance amplifier to select the resonant frequency.The frequency controller may be further adapted to modify a currentthrough the negative transconductance amplifier in response to a voltageor in response to fabrication process variation. The frequencycontroller may be further adapted to modify a transconductance of thenegative transconductance amplifier in response to fabrication processvariation.

The frequency controller may also include a voltage isolator coupled tothe resonator and adapted to substantially isolate the resonator from avoltage variation, such as a current mirror, or a current mirror whichhas a cascode configuration.

In another exemplary embodiment, the present invention provides anapparatus, comprising a resonator adapted to provide a first signalhaving a resonant frequency; a negative transconductance amplifiercoupled to the resonator; and a temperature compensator coupled to thenegative transconductance amplifier and to the resonator, thetemperature compensator adapted to modify the resonant frequency inresponse to temperature.

The temperature compensator is further adapted to modify a currentthrough the negative transconductance amplifier in response totemperature, and may further comprise a current source responsive totemperature. The current source may have one or more configurationsselected from a plurality of configurations, the plurality ofconfigurations comprising CTAT, PTAT, and PTAT² configurations. Thecurrent source is typically coupled though one or more current mirrorsto the negative transconductance amplifier.

The current source may include a first transistor; a second transistorcoupled to the first transistor; a diode coupled to the firsttransistor; and a resistor coupled to the second transistor. The currentprovided by the current source is a function of a voltage across thediode and a resistance of the resistor, wherein the voltage and theresistance are temperature-dependent, and wherein the first and secondtransistors are operable in strong inversion.

Another, second current source may include a first transistor; a secondtransistor coupled to the first transistor; and a resistor coupled tothe second transistor. The current provided by the second current sourceis a function of a voltage across the resistor, a resistance of theresistor, and respective sizes of the first and second transistor,wherein the voltage and the resistance are temperature-dependent, andwherein the first and second transistors are operable at a subthresholdvoltage.

Another, third current source may include a plurality of transistors;and a resistor coupled to a transistor of the plurality of transistors.The current provided by the third current source is a function of asquare of a voltage across the resistor, wherein the voltage istemperature-dependent. In the third current source, a first set oftransistors of the plurality of transistors are operable in stronginversion and a second set of transistors of the plurality oftransistors are operable at a subthreshold voltage.

In another exemplary embodiment, the invention provides an apparatuscomprising a resonator, the resonator adapted to provide a first signalhaving a resonant frequency; a negative transconductance amplifiercoupled to the resonator; a current mirror coupled to the negativetransconductance amplifier; and a current source coupled to the currentmirror, the current source adapted to modify the resonant frequency byvarying a current through the current mirror and the negativetransconductance amplifier in response to temperature. The currentsource may have one or more configurations selected from a plurality ofconfigurations, the plurality of configurations comprising CTAT, PTAT,and PTAT² configurations. The apparatus may also include a plurality ofcurrent sources coupled to the current mirror, with the a plurality ofcurrent sources having at least two configurations selected from aplurality of configurations, the plurality of configurations comprisingCTAT, PTAT, and PTAT² configurations.

Also in summary, the present invention provides an apparatus comprisinga resonator adapted to provide a first signal having a resonantfrequency; an amplifier coupled to the resonator; and a frequencycontroller (coupled to the resonator) which is adapted to select aresonant frequency having a first frequency of a plurality offrequencies. The apparatus also includes a frequency divider (coupled tothe resonator) which is adapted to divide the first signal having thefirst frequency into a plurality of second signals having acorresponding plurality of frequencies, the plurality of frequenciessubstantially equal to or lower than the first frequency, such as bydivision by a rational number.

The first signal may be a differential signal or a single-ended signal.When the first signal is a differential signal, the frequency divider isfurther adapted to convert the differential signal to a single-endedsignal. Similarly, when the first signal is a substantially sinusoidalsignal, the frequency divider is further adapted to convert thesubstantially sinusoidal signal to a substantially square wave signal.

In various embodiments, the frequency divider may comprise a pluralityof flip-flops or counters coupled successively in series, wherein anoutput of a selected flip-flop or counter is a frequency of a previousflip-flop or counter divided by two, or more generally, a plurality ofdividers coupled successively in series, wherein an output of asuccessive divider is a lower frequency than the output of a previousdivider. The plurality of dividers may be differential, single-ended, ordifferential and single-ended, such as differential followed by a finalsingle-ended stage. The frequency divider may also include a square-wavegenerator adapted to convert the first signal into a substantiallysquare-wave signal having a substantially equal high and low duty cycle.

The present invention may also include a frequency selector coupled tothe frequency divider, and adapted to provide an output signal from theplurality of second signals. The frequency selector may further comprisea multiplexer and a glitch-suppressor.

The present invention may also include a mode selector coupled to thefrequency selector, wherein the mode selector is adapted to provide aplurality of operating modes, which may be selected from a groupcomprising a clock mode, a timing and frequency reference mode, a powerconservation mode, and a pulse mode.

For a reference mode, the invention may also include a synchronizationcircuit coupled to the mode selector; and a controlled oscillatorcoupled to the synchronization circuit and adapted to provide a thirdsignal; wherein in the timing and reference mode, the mode selector isfurther adapted to couple the output signal to the synchronizationcircuit to control timing and frequency of the third signal. Such asynchronization circuit may be a delay-locked loop, a phase-locked loop,or an injection locking circuit.

In selected embodiments, the amplifier may be a negativetransconductance amplifier. The frequency controller may be furtheradapted to modify a current through the negative transconductanceamplifier in response to temperature, and may comprise a current sourceresponsive to temperature. Such a current source may have one or moreconfigurations selected from a plurality of configurations, such as theplurality of configurations comprising CTAT, PTAT, and PTAT²configurations. In addition, the frequency controller may be furtheradapted to modify a current through the negative transconductanceamplifier to select the resonant frequency, modify a transconductance ofthe negative transconductance amplifier to select the resonantfrequency, or modify a current through the negative transconductanceamplifier in response to a voltage. The frequency controller may alsoinclude a voltage isolator coupled to the resonator and adapted tosubstantially isolate the resonator from a voltage variation, and maycomprises a current mirror, which may further have a cascodeconfiguration. The frequency controller may be further adapted to modifya capacitance or an inductance of the resonator in response tofabrication process variation, temperature variation, or voltagevariation.

The frequency controller may have various embodiments for these variousfunctions, and may further comprise: a coefficient register adapted tostore a first plurality of coefficients; and a first array having aplurality of switchable capacitive modules coupled to the coefficientregister and to the resonator, each switchable capacitive module havinga fixed capacitance and a variable capacitance, each switchablecapacitive module responsive to a corresponding coefficient of the firstplurality of coefficients to switch between the fixed capacitance andthe variable capacitance and to switch each variable capacitance to acontrol voltage. The plurality of switchable capacitive modules may bebinary-weighted, or have another weighting scheme. The frequencycontroller may also include a second array having a plurality ofswitchable resistive modules coupled to the coefficient register andfurther having a capacitive module, the capacitive module and theplurality of switchable resistive modules further coupled to a node toprovide the control voltage, each switchable resistive module responsiveto a corresponding coefficient of a second plurality of coefficientsstored in the coefficient register to switch the switchable resistivemodule to the control voltage node; and a temperature-dependent currentsource coupled through a current mirror to the second array.

The frequency controller may also include a process variationcompensator coupled to the resonator and adapted to modify the resonantfrequency in response to fabrication process variation. In an exemplaryembodiment, the process variation compensator may comprise: acoefficient register adapted to store a plurality of coefficients; andan array having a plurality of switchable capacitive modules coupled tothe coefficient register and to the resonator, each switchablecapacitive module having a first fixed capacitance and a second fixedcapacitance, each switchable capacitive module responsive to acorresponding coefficient of the plurality of coefficients to switchbetween the first fixed capacitance and the second fixed capacitance.The plurality of switchable capacitive modules may be binary-weighted,or have another weighting scheme.

In another exemplary embodiment the process variation compensator maycomprise: a coefficient register adapted to store a plurality ofcoefficients; and an array having a plurality of switchable variablecapacitive modules coupled to the coefficient register and to theresonator, each switchable variable capacitive module responsive to acorresponding coefficient of the plurality of coefficients to switchbetween a first voltage and a second voltage. The plurality ofswitchable variable capacitive modules also may be binary-weighted, orhave another weighting scheme.

The present invention may also include a frequency calibration modulecoupled to the frequency controller and adapted to modify the resonantfrequency in response to a reference signal. For example, the frequencycalibration module may include a frequency divider coupled to thefrequency controller, the frequency divider adapted to convert an outputsignal derived from the first signal having the first frequency to alower frequency to provide a divided signal; a frequency detectorcoupled to the frequency divider, the frequency detector adapted tocompare the reference signal to the divided signal and provide one ormore up signals or down signals; and a pulse counter coupled to thefrequency detector, the pulse counter adapted to determine a differencebetween the one or more up signals or down signals as an indicator of adifference between the output signal and the reference signal.

The resonator used with the invention may comprise an inductor (L) and acapacitor (C) coupled to form an LC-tank, having a selectedconfiguration of a plurality of LC-tank configurations, such as series,parallel and so on, and may include other components. In otherembodiments, the resonator may be selected from a group comprising: aceramic resonator, a mechanical resonator, a microelectromechanicalresonator, and a film bulk acoustic resonator, or any other resonatorwhich is electrically equivalent to an inductor (L) coupled to acapacitor (C).

The apparatus of the invention may be utilized as a timing and frequencyreference, or as a clock generator. In addition, the invention may alsoinclude a second oscillator (such as a ring, relaxation, or phase shiftoscillator) providing a second oscillator output signal; and a modeselector coupled to the frequency controller and to the secondoscillator, the mode selector adapted to switch to the second oscillatoroutput signal to provide a power conservation mode. Additional operatingmodes may be provided by a mode selector coupled to the frequencycontroller, which may be adapted to periodically start and stop theresonator to provide a pulsed output signal, or adapted to selectivelystart and stop the resonator to provide a power conservation mode.

In another selected embodiment, the apparatus of the invention,comprises: a resonator adapted to provide a first signal having aresonant frequency; an amplifier coupled to the resonator; a temperaturecompensator coupled to the amplifier and to the resonator, thetemperature compensator adapted to modify the resonant frequency inresponse to temperature; a process variation compensator coupled to theresonator, the process variation compensator adapted to modify theresonant frequency in response to fabrication process variation; afrequency divider coupled to the resonator, the frequency divideradapted to divide the first signal having the resonant frequency into aplurality of second signals having a corresponding plurality offrequencies, the plurality of frequencies substantially equal to orlower than the resonant frequency; and a frequency selector coupled tothe frequency divider, the frequency selector adapted to provide anoutput signal from the plurality of second signals.

In another selected embodiment, the apparatus of the invention generatesa clock signal, and comprises: an LC resonator adapted to provide adifferential, substantially sinusoidal first signal having a resonantfrequency; a negative transconductance amplifier coupled to the LCresonator; a temperature compensator coupled to the negativetransconductance amplifier and to the LC resonator, the temperaturecompensator adapted to modify a current in the negative transconductanceamplifier in response to temperature and further to modify a capacitanceof the LC resonator in response to temperature; a process variationcompensator coupled to the LC resonator, the process variationcompensator adapted to modify the capacitance of the LC resonator inresponse to fabrication process variation; a frequency divider coupledto the resonator, the frequency divider adapted to convert and dividethe first signal having the resonant frequency into a plurality ofsingle-ended, substantially square-wave second signals having acorresponding plurality of frequencies, the plurality of frequenciessubstantially equal to or lower than the resonant frequency, and eachsecond signal having a substantially equal high and low duty cycle; anda frequency selector coupled to the frequency divider, the frequencyselector adapted to provide an output signal from the plurality ofsecond signals.

From the foregoing, it will be observed that numerous variations andmodifications may be effected without departing from the spirit andscope of the novel concept of the invention. It is to be understood thatno limitation with respect to the specific methods and apparatusillustrated herein is intended or should be inferred. It is, of course,intended to cover by the appended claims all such modifications as fallwithin the scope of the claims.

1. An apparatus for frequency control of a reference signal; theapparatus comprising: a reference resonator adapted to provide thereference signal having a resonant frequency; a negativetransconductance amplifier coupled to the reference resonator; and afrequency controller coupled to the negative transconductance amplifierand coupled to the reference resonator, the frequency controller adaptedto modify the resonant frequency in response to at least one variable ofa plurality of variables.
 2. The apparatus of claim 1, wherein theplurality of variables comprise temperature, fabrication process,voltage, and frequency.
 3. The apparatus of claim 1, wherein thefrequency controller is further adapted to modify a current through thenegative transconductance amplifier in response to temperature.
 4. Theapparatus of claim 3, wherein the frequency controller further comprisesa current source responsive to temperature.
 5. The apparatus of claim 4,wherein the current source has one or more configurations selected froma plurality of configurations, the plurality of configurationscomprising CTAT, PTAT, and PTAT² configurations.
 6. The apparatus ofclaim 1, wherein the frequency controller is further adapted to modify acurrent through the negative transconductance amplifier to select theresonant frequency.
 7. The apparatus of claim, 1, wherein the frequencycontroller is further adapted to modify a transconductance of thenegative transconductance amplifier to select the resonant frequency. 8.The apparatus of claim 1, wherein the frequency controller is furtheradapted to modify a current through the negative transconductanceamplifier in response to a voltage.
 9. The apparatus of claim 1, whereinthe frequency controller is further adapted to modify a transconductanceof the negative transconductance amplifier in response to fabricationprocess variation.
 10. The apparatus of claim 1, wherein the frequencycontroller is further adapted to modify a current through the negativetransconductance amplifier in response to fabrication process variation.11. The apparatus of claim 1, wherein the frequency controller furthercomprises a voltage isolator coupled to the resonator and adapted tosubstantially isolate the resonator from a voltage variation.
 12. Theapparatus of claim 11, wherein the voltage isolator comprises a currentmirror.
 13. The apparatus of claim 12, wherein the current mirror has acascode configuration.
 14. The apparatus of claim 1, wherein theresonator is one or more of the following resonators: an inductor (L)and a capacitor (C) configured to form an LC-tank resonator; a ceramicresonator, a mechanical resonator, a microelectromechanical resonator,and a film bulk acoustic resonator.
 15. An apparatus, comprising: areference resonator, the reference resonator adapted to provide areference signal having a resonant frequency; a negativetransconductance amplifier coupled to the reference resonator; and atemperature compensator coupled to the negative transconductanceamplifier and to the reference resonator, the temperature compensatoradapted to modify the resonant frequency in response to temperature. 16.The apparatus of claim 15, wherein the temperature compensator isfurther adapted to modify a current through the negativetransconductance amplifier in response to temperature.
 17. The apparatusof claim 16, wherein the temperature compensator further comprises acurrent source responsive to temperature.
 18. The apparatus of claim 17,wherein the current source further comprises: a first transistor; asecond transistor coupled to the first transistor; a diode coupled tothe first transistor; and s a resistor coupled to the second transistor.19. The apparatus of claim 18, wherein the current provided by thecurrent source is a function of a voltage across the diode and aresistance of the resistor, wherein the voltage and the resistance aretemperature-dependent.
 20. The apparatus of claim 18, wherein the firstand second transistors are operable in strong inversion.
 21. Theapparatus of claim 17, wherein the current source further comprises: afirst transistor; a second transistor coupled to the first transistor;and a resistor coupled to the second transistor.
 22. The apparatus ofclaim 21, wherein the current provided by the current source is afunction of a voltage across the resistor, a resistance of the resistor,and respective sizes of the first and second transistor, wherein thevoltage and the resistance are temperature-dependent.
 23. The apparatusof claim 21, wherein the first and second transistors are operable at asubthreshold voltage.
 24. The apparatus of claim 17, wherein the currentsource further comprises: a plurality of transistors; and a resistorcoupled to a transistor of the plurality of transistors.
 25. Theapparatus of claim 24, wherein the current provided by the currentsource is a function of a square of a voltage across the resistor,wherein the voltage is temperature-dependent.
 26. The apparatus of claim24, wherein a first set of transistors of the plurality of transistorsare operable in strong inversion and a second set of transistors of theplurality of transistors are operable at a subthreshold voltage.
 27. Theapparatus of claim 17, wherein the current source has one or moreconfigurations selected from a plurality of configurations, theplurality of configurations comprising CTAT, PTAT, and PTAT²configurations.
 28. The apparatus of claim 17, wherein the currentsource is coupled though one or more current mirrors to the negativetransconductance amplifier.
 29. An apparatus, comprising: a referenceresonator, the reference resonator adapted to provide a reference signalhaving a resonant frequency; a negative transconductance amplifiercoupled to the reference resonator; a current mirror coupled to thenegative transconductance amplifier; and a current source coupled to thecurrent mirror, the current source adapted to modify the resonantfrequency by varying a current through the current mirror and thenegative transconductance amplifier in response to temperature.
 30. Theapparatus of claim 29, wherein the current source has one or moreconfigurations selected from a plurality of configurations, theplurality of configurations comprising CTAT, PTAT, and PTAT²configurations.
 31. The apparatus of claim 30, further comprising aplurality of current sources coupled to the current mirror, theplurality of current sources having at least two configurations selectedfrom a plurality of configurations, the plurality of configurationscomprising CTAT, PTAT, and PTAT² configurations.
 32. An apparatus,comprising: a resonator, the resonator adapted to provide a first signalhaving a resonant frequency; a negative transconductance amplifiercoupled to the resonator; a current minor coupled to the negativetransconductance amplifier; and a plurality of current sources coupledto the current mirror, the plurality of current sources adapted tomodify the resonant frequency by varying a current through the currentmirror and the negative transconductance amplifier in response totemperature, the plurality of current sources having at least twoconfigurations selected from a plurality of configurations comprisingCTAT, PTAT, and PTAT² configurations.